The invention concerns a frequency modulated converter with a series-parallel resonance, particularly for driving any ohmic or inductive load, including gas discharge tubes, wherein a commutating voltage switch in the form of a transistor is provided in is connected in series between the negative electrode of a direct current voltage source and a first terminal of an inductor, wherein a pulse generator circuit is provided between the voltage source and a control electrode of the transistor and wherein a second terminal of the inductor is connected to the primary winding of a transformer.
The invention further comprises a first capacitor and a rectifier diode provided in a first and second parallel branch respectively between the charge emitting and the charge receiving electrode of the transistor, and a second capacitor is provided across the electrodes of the voltage source and additionally provides a smoothing capacitance for the voltage source, the second capacitor being connected in series with the inductor via the diode.
The later years have seen a dramatic reduction in the physical dimensions of power converters, something which has been achieved by increasing the operating frequencies. Common quasi-square pulse converters now approach an operating range whose upper limit lies about 0.5 MHz. This allows a significant reduction in the size of the most important passive power components such as the magnetic components and the capacitors, compared to for instance with switches for 20 kHz. However, these converters, being for the greater part pulse width modulated converters, have high switching losses in the power semiconductors, something which leads to reduced efficency and hence a need for more cooling, which decreases the possibilities of a reduction in the physical dimensions of the converters.
In order to increase the capability of power converters to handle overloads or large load varations, for instance when used as power supplies for gas discharge tubes, there has been proposed to monitor or avoid saturation of the switching transistors by special circuit arrangements, as will be evident from for instance PCT application NO No. WO 90/01248 and GB-PS-No. 1 378 465.
A more effective way of converting power at increasingly higher frequencies is based on the so-called xe2x80x9czero current switchingxe2x80x9d wherein a sinusoidal voltage which may be generated by a LC-resonant tank connected either in parallel or series, is used. Such converters are called xe2x80x9cresonant convertersxe2x80x9d. The advantage of using a sine voltage is that losses in the power semiconductors are dramatically reduced, as the switching generally takes place at the zero crossing. The disadvantage of resonant converters is that at a given power level the peak current is many times greater than that of a pulse width modulated converter. By using semiconductors with lower resistance in conduction it is however possible to increase the operating frequencies to above 1 MHz. Thus power densities well beyond 1 W/cm3 may be attained.
For use in such converters there is now known a frequency modulated control in the form of an integrated circuit which may be used in the range beyond 1 MHz, with the designation LD 405, obtainable from Gennum Corporation, Burlington, Ontario, Canada. The use of this control circuit in a frequency modulated converter is described in a LD 405 application note from Gennum Corporation with the title xe2x80x9cUsing LD 405 in a 125 W resonant mode power supplyxe2x80x9d. To this end the same corporation has introduced a resonant circuit of which the embodiment in principle is shown in the appended FIG. 1. The circuit comprises an inductance L, a capacitance C, a resistance R and a load RL. Before the inductance L a commutating switch S, for instance in the form of a transistor, has been provided its purpose is to supply direct current from a source V to a series resonant tank LC. The resistance of the load RL drains current from the tank. As soon as the resonant process has been terminated, the switch S opens and the power conversion from the source S to the load RL is interrupted. After a given period of Lime time the switch S again closes and the process is repeated. The commutation frequency may be changed such that the average power dissipated in the load RL is changed.
In a practical embodiment a resonant converter of this type works with two commutating switches each of which handles a respective half-cycle of the resonant period. The switches are based on MOS field effect transistors which themselves are driven by a respective MOSFET stage. The output stage of the shown embodiment is based on Schottky rectifier diodes.
However, with this prior art embodiment of the resonant converter it is difficult wholly to avoid harmonics in the resonant voltage and it is also difficult to symmetrisize the half-cycles such that they get the same energy content. Finally there will still be substantial losses in the power switches and the Schottky output diodes. Beyond this RC networks have been provided across the power switches in order to dampen voltage transients and these damping circuits lead to additional losses. Hence the efficiency is reduced by at least 25% and even if an output stage without rectifying diodes is used, the loss will be around 16%.
Generally it can be said of converters as discussed above and prior art devices of the same type that the capacitor is connected in direct parallel with the inductor and the switch in series with the voltage supply. In addition the load will steal energy from the resonant circuit which also may be made as a transformer. These prior art devices are generally difficult to compute and realize due to the limitation in the energy content of the resonance. If too much energy is drained from the LC circuit, the frequency changes and it is necessary with complicated electronic control means in order to control the switching of the commutating switches such that the resonant state of the circuit is maintained. If overload arises in such circuits, the switching current of the transistor increases uncontrollably and if the transistors are disconnected, this may lead to transients which incur irrepearable irreparable damage to the converter. The problem is that the control means which is to protect the transistor, does not work in real time and hence the transistor, i.e. the switch, is exposed to non-normal loads. As already mentioned the substantial losses are still present, such that the efficiency of the converter does not attain more than about 84% without the use of a rectified output.
Finally U.S. Pat. No. 4,613,769 discloses a transistor oscillator circuit wherein a capacitor is connected in parallel over the terminals of a secondary winding in a transformer, the latter operating in parallel sine wave resonance with the capacitor as a first wave-shaping means at a given frequency. A second wave-shaping means consists of another capacitor connected in parallel over the collector and emitter electrodes of the transistor oscillator and operates in series resonance with an inductor, at twice the given frequency.
The object of the present invention is to provide a resonant circuit without the above-mentioned and other disadvantages. This is achieved according to the invention in that the first capacitor and the inductor together form a series resonance circuit, the relationship between the inductor voltage UL and the capacitance of the first capacitor determining the series resonance frequency of a first half-cycle, that the second capacitor and the diode together form a parallel resonance circuit, the inductor voltage UL and the capacitance of a second capacitor determining the parallel resonance frequency of a second half-cycle, that the transistor is in a high ohmic state during both the series and parallel resonance mode, that the diode acting as an impedance selector between the capacitors to maintain the correct current flow in the transformer and the load is conducting in the parallel resonance mode, charging the capacitor above the voltage source level, before the transistor is switched into a low ohmic state and completes that parallel resonance mode and then initiates another series-parallel resonance when switched into the high ohmic state, each half-cycle of the resonant period being kept in time of the switching of the transistor into the high ohmic state, the transformer, the inductor and the capacitors thus constituting an RCL resonator operating in series-parallel to the transistor, the quality factor of the resonant being determined by the relationship between the inductor voltage UL and the capacitor impedances ZC1 and ZC2 respectively and the supply voltage UL, and that the load is connected between the terminals of a first secondary winding in the transformer, such that the load is connected in series with the inductor consuming energy in each half cycle of the resonance period from both the inductor and the direct voltage source, the transistor thus operating as a commutating voltage switch in series with the voltage source in the first half-cycle and in parallel with the voltage source in the second half-cycle, all the time carrying a fraction of the total energy consumed by the load.
To these ends, the present invention also consists of a frequency-modulated converter with series-parallel resonance, particularly for driving any ohmic or inductive load (RG), including gas discharge tubes, wherein a commutating voltage switch (Q) in the form of a transistor is connected in series between the negative electrode of a direct voltage source and a first terminal of an inductor (L,P), wherein a pulse generator circuit is provided between the voltage source and a control electrode of the transistor (Q) with a transformer (T), and further comprises: a first capacitor (C1) and a rectifier diode (D2) provided in a first and second parallel branch respectively between the charge emitting and the charge receiving electrode of the transistor (Q); a second capacitor (C3) provided across the electrodes of the voltage source, the second capacitor (C3) being connected in series with the inductor (L) via the diode (D2); the first capacitor (C1) and the inductor (L) together forming a series resonance circuit, the relationship between the inductor voltage UL and the capacitance of the first capacitor (C3) determining the series resonance frequency of a first half-cycle; the second capacitor (C3) and the inductor L together forming a parallel resonance circuit, the relationship between inductor voltage UL and the capacitance of the second capacitor (C3) determining the parallel resonance frequency of a second half-cycle, the capacitor (C3) having a capacitance which is several times greater than that of the capacitor (C1), that the transistor (Q) is in a high ohmic state during both the series and parallel resonance mode; the diode (D2) acting as an impedance selector between the capacitors (C1, C3) to maintain the correct current flow in the transformer (T) and the load (RG) is conducting in the parallel resonance mode, charging the capacitor (C3) above the voltage source level, before the transistor (Q)is switched into a low ohmic state and completes the parallel resonance mode and then initiates another series-parallel resonance when switched into the high ohmic state, each half-cycle of the resonant period being kept in time by the switching of the transistor (Q) into the high ohmic state, the transformer (T), the inductor (L), and the capacitors (C1, C3) thus constituting an RCL resonator operating in series-parallel to the transistor (Q) voltage source, the quality factor of the resonator being determined by the relationship between the inductor voltage UL and the capacitor impedances ZC1 and ZC3 respectively and the supply voltage U, and the load (RG) being connected between the terminals of a first secondary winding (S1) in the transformer (T), such that the load (RG) is connected in series with the inductor (L,P) consuming energy in each half-cycle of the resonance period from both the conductor and the direct voltage source, the transistor (Q) thus operating as a commutating voltage switch in series with the voltage source in the first half-cycle and in parallel with the voltage source in the second half-cycle, all the time carrying a fraction of the total energy consumed by the load (RG). Further features and advantages are evident from the appended dependent claims.
The invention shall now be discussed in greater detail below with reference to the appended drawing.